Motor Lead Length Issues for IGBT PWM Drives
As PWM variable frequency technology advances in the use of IGBT power transistors, concerns have arisen over the amount of conductor located between the controller and the low voltage induction motor. Several technical papers have been presented on the subject of dv/dt from these controllers and the effects on motor insulation. This paper will build on these previous papers while detailing application of transmission theory, drive current feedback design impact, proper modeling techniques of conductor distributed impedance, and motor design considerations. Theoretical and Experimental test data are included to support the findings. Although the focus of this paper is on applications below 5 HP, some theories and tests presented should be considered in larger systems. Chipping, De-Barking, Washing, and Coating applications in the industry sometimes do not lend themselves well to close proximity location of motors and controllers. IGBT PWM design, motor design, and installation guidelines will be discussed as solutions with several options presented to the engineer.
Insulated Gate Bipolar Transistor (IGBT) technology has quickly been gaining popularity among the PWM AC inverter applications. In particular, the usage of IGBTs dramatically increased for PWM Drives rated up to 200 horsepower in place of the traditional bipolar transistor. Performance benefit is the motivation behind this trend which are characterized as a low acoustic motor noise due to high switching frequency, low torque ripple, and low losses associated with the switching device superiority.
At the same time, this relatively new inverter technology is creating new problems in its application. The motor lead length combined with the high switching frequency becomes a more sensitive issue for proper drive operation. Effects of the long lead motor cable and high switching frequency should be emphasized for both motor and inverter.
For the motors, IGBT PWM Drives can sometimes create insulation breakdown between the phase windings due to high transient voltage peaking. For the drive, nuisance overcurrent trips can result due to a high frequency oscillation of the current waveforms.
Focus needs to be placed on several factors. Voltage Standing Waves, capacitive coupling between wires, the distributed impedance model of wire, the dv/dt of the PWM switching, and sensitivity of the current feedback circuit in the inverter are of our most concerns. This paper describes the phenomenon, the experiment results, and the theoretical analysis based on transmission theory of the motor lead length combined with the recent IGBT high switching inverter technology.
II. IGBT Transistor Characteristics - Background Information
The IGBT is basically a combination of a Metal Oxide Silicon Field Effect Transistors (MOSFET) and a bipolar transistor. At the switching frequencies of current PWM motor drives, IGBT's can carry more current than a similar sized bipolar transistor, and they require lower control power like a MOSFET. Also the on-state voltage of an IGBT is similar to the on-state voltage of a Darlington bipolar transistor, and IGBTs are capable of switching at a higher frequency. These characteristics make the IGBT the best choice for the power switching device in low voltage PWM motor drives.
The structure of an IGBT die is similar to a N channel MOSFET, with one added junction, a Player on the bottom of the device. This added junction effectively becomes the collector of the PNP bipolar transistor, which is driven by the N channel MOSFET. This structure takes advantage of conductivity modulation to reduce the on-state voltage of the device, as compared to a similar sized MOSFET. The extra junction also increases the blocking voltage in the off-state. One disadvantage of the IGBT as compared to a MOSFET is that the conductivity modulation limits the switching speed. Even so the IGBT is still faster than a bipolar Darlington transistor.
IGBTs can be optimized for either switching speed or on-state voltage. These two parameters oppose each other. In order to make the devices switch faster, various doping, irradiation, and minority carrier lifetime control techniques are used, which increase the on-state voltage. Most IGBT manufacturers make 2 or 3 classes of devices, some optimized for speed and others optimized for on-state voltage. For a particular PWM motor drive, the switching frequency must be considered when selecting which IGBT to use.
IGBTs have evolved and improved over the past few years. The first generation IGBTs had problems such as latch up, long current tail time, and low SOA. These problems have been eliminated in the latest generation devices. Also the on-state voltage and switching speed have been improved, so that IGBTs are approaching the theoretical limits on these characteristics. Most manufacturers can make a device today which has an on-state voltage that is a factor of 2 lower than that of earlier generation devices, while still maintaining the same switching speed.
AC drives using IGBTs almost always include a free-wheeling diode in parallel with each IGBT in the package. These diodes have also been improved recently. In earlier generation modules these diodes were "snappy", ie. the reverse recovery current changed at a high rate, causing a large voltage spike through the parasitic inductances in the module. Now most manufacturers use soft recovery diodes, which turn off smoothly. These new diodes are also optimized to reduce the module switching losses, by doping and irradiation to reduce the total reverse recovery time. Although this increases their on-state voltage, it reduces the loss in the IGBT, and the total loss of the module is lower.
Safe Operation Area (SOA) of IGBTs, during both turn-off and short circuit, are very important characteristics for motor drives. With the inductive load from a motor drive, when the IGBT turns off the voltage rises before the current falls. During this time the voltage and current must remain within the RBSOA (or switching SOA) rating of the IGBT. The other SOA rating is important because a motor drive must survive a short circuit on the output terminals. This is when the Short Circuit SOA becomes important. During this time the IGBT has high voltage across it, and much greater than rated current at the same time. The power dissipation in the device at this time is magnituded greater than the device can withstand continuously. This condition must be sensed, and the IGBT must be turned off within a specified period of time, usually 10 micro seconds.
There are many PWM methods used in the AC PWM motor drives. The design choice depends on the primary functional objective of the drive. For example, if the voltage utilization and the output voltage capability are of primary concern, either space vector modulation or harmonic injection with overmodulation is a very good method. These effective harmonic injection and overmodulation techniques are in fact very common among IGBT inverters.
The switching frequency at the motor terminal varies depending on the modulation index, the modulation scheme, the dead time, etc. When the modulation index exceeds unity (i.e. overmodulation region), the switching frequency is reduced due to PWM pulse concentration over the PWM cycles. When the modulation index becomes near zero, the switching frequency is also reduced due to the dead time effect. The switching frequency becomes maximum at a modulation index equal to .5.
This implies that both motor and inverter become most susceptible at around 30Hz output frequency assuming a 60Hz base frequency.
IV. Analysis based on Transmission Line Theory
The fast switching of PWM drives create significant transient surges that can overstress the motor winding insulation systems, the inverter protection systems, and the various terminal connections within the drive system. To analyze these phenomenon, classical transmission line theory can be applied to understand the voltage and current reflection at the motor and drive terminals, and the distributed motor winding model can be used to identify the intra-winding voltage stress. Considering only the terminal characteristics, the voltage standing wave ratio, VSWR, and the reflection coefficients, rv, can be defined as follows:
Where V1 and V0 represent the reflected and incident voltage seen at the terminal of the AC motor.
The machine characteristic impedance and the cable characteristic impedance are represented by ZL and Z0, respectively. Since the motor is essentially an inductive apparatus, ZL can be more than ten times Z0. As a result, the reflected voltage magnitude is typically between 80% and 100% of that of the incident voltage . The cable characteristic impedance can be approximated by the two-wire transmission line given by:
Where 'D' is the distance between lines, and 'a' is the conductor diameter. For closely coupled cables, it typically varies between 120 and 150 ohms.
An unequal relationship between motor and cable characteristic impedance can be found in relatively low HP motors. For example, a 1HP AC motor characteristic impedance is typically ten times larger than that of the cable characteristic impedance. Because of this mismatch of the characteristic impedance, the voltage standing wave becomes a significant issue when the motor is driven by high switching frequency voltage through long lead cables.
The total voltage at the motor terminal can be written as the vector summation of V1 and V0. The phase velocity of the traveling waves can be calculated from the distributed cable inductance and capacitance . For a given cable length, the phase velocity of the traveling waves provides propagation delay time, tp, which can be more than twice as long as the voltage rise time due to the fast IGBT switching. Under such conditions, the incident voltage and reflected waves at the motor terminal are almost in phase, and the motor terminal voltage can reach almost twice the rated value by the scaler summation.
To illustrate this point, simulation
and experimental tests were conducted.
A PSPICE simulation was performed to
evaluate the effect of the long lead cable. Figure 1
shows the circuit for the simulation, which consists of
the IGBT inverter bridge, the lossy transmission model of
the 250 feet cable, and the AC motor. The lossy
transmission line has a lumped sum model consisting of a
0.166 ohms series resistance and 50 uH series inductance
and 1000PF capacitance between phases. These values are
actually measured values of the 250 feet No.12 AWG size
cables. The PWM scheme is based on plain sinusoidal and
asynchronous modulation. The operating conditions are
based on a 8kHz carrier, 650V DC bus voltage which is
equivalent to 460V AC input voltage to the inverter, and
60Hz output frequency.
Figure 2 shows the line-to-line voltage
over four carrier frequency switching cycles at the motor
connection point. At every switching transitions, the
surge voltage reaches maximum 2pu with 400kHz frequency
oscillation due to a collision between the incident and
reflected voltage waves. This high frequency voltage
oscillation creates a harmful current flow due to the
distributed capacitance coupling across the phase lines.
The simulated current waveform at the inverter connection
point is shown in Figure 3, which also contains the same
frequency oscillation synchronous to the voltage
This experimental test was performed by
using an open loop V/Hz IGBT inverter with a 8kHz carrier
frequency. The four different motor lead lengths, 250,
500, 750, and 1000 feet, were tested. The motor leads
used were four No.12 AWG wires containing three phase
wires and the ground wire which are closely coupled
together inside a one inch diameter aluminum conduit. The
AC induction motors tested was a 1HP/460V rated at 1.5A
and 10HP/460V rated at 12.6A.
The voltage and current waveforms were
measured by an oscilloscope at both motor and inverter
ends. Figure 4 and 5 show the measured line-to-line
voltage of 1HP/460V drive at both the inverter and motor
ends with a 250 foot length cable. The upper trace is the
voltage at the motor end and the lower trace is the
voltage at the inverter end. Both traces are 400V/DIV.
Figure 6 and 7 are also the voltage waveforms of the
1HP/460V drive except the cable length is 500 feet.
It is clear that the ringing frequency
changes from 425kHz to 238kHz as the cable length
increases from 250 feet to 500 feet.
Figure 8 and 9 are the current waveforms for both 1HP/460V and 10HP/460V drives with 250feet cable length. Figure 8 shows the current waveforms of the 1HP/460V drive under the full load condition (1.5A) at both inverter (upper trace) and motor ends (lower trace). The current waveforms at the inverter end contains the high frequency ringing oscillation. This oscillation frequency is the same frequency in the voltage waveform at the motor end. Therefore, the frequency will increase as the cable length increases.
Figure 9 shows the inverter end current waveforms at a full load condition with the same 250 feet output cable length.
When making a comparison between two drive current waveforms, the 1HP/460V actually created a nuisance overcurrent trip due to the relatively large and high frequency oscillation detected by the inverter overcurrent trip circuit. The 10HP/460V drive successfully ran without any nuisance trips.
Characteristic impedance mismatch between the motor and the cable is greater in the 1HP motor than that in the 10HP motor. The 10HP motor has smaller characteristic impedance and is relatively close by comparison to the impedance of the cable.
The reflection coefficients and VSWR were also measured by the network analyzer, HP3589A.
Figure 10 through 11 show the Smith
Chart for 250 feet, and 1000 feet lead length. In each
Figure, the cursor is located where the ringing frequency
occurred at each switching in the time domain. All
Figures show consistent location on the Smith Chart which
corresponds to r = -73ƒ ~ -82ƒ. This clearly indicates that
the load is capacitive, and VSWR is a very high value,
which is approximately between 5 ~ 20, and
correspondingly |r| = 0.67 ~ 0.90. Also, the load impedance of
each lead length cables is found from the Smith Chart
such as 0.1-j1.2 for 250 feet cable, and 0.3-j1 for 1000
feet cable, respectively. As the lead length increases,
the real part of the load impedance also increases due to
the increasing resistance.
For the motor winding surges, a distributed winding model must be considered. In this case the winding can be represented as in Figure 12. Depending on the connections of the windings, the initial voltage distribution (t=0) can be derived as follows.
The final voltage
distributions for the above two cases can also be derived
represents the square root of the ratio between
the inter-terminal capacitance and the winding
capacitance to ground, x the particular
displacement in the winding and l the overall
length of the winding.
The difference between the final and initial voltage distributions represent the magnitude of the voltage resonance within the winding. The higher the difference, the worse the resonance condition. Similarly, with a higher voltage gradient between the initial and final voltage distribution, the intra-winding gradient voltage as well as winding-to-ground voltage can all be much larger.
The high frequency oscillation of the
current waveforms makes design a very difficult trade off
between the filtering and the overcurrent protection
circuit in the inverter. The proper filtering is required
to extract the fundamental frequency component of the
current waveforms. Particularly precise current feedback
information is essential for those drives which
incorporate the closed loop Vector current control. On
the other hand, the recent IGBTs mentioned earlier
require less than 10 usec short circuit pulsation on the
non repetitive basis. Typical short circuit protection
delay is chosen around 2 ~ 5 microseconds, which
corresponds to a 200 ~ 500kHz filter cut-off frequency.
Today, most IGBT based inverters use very fast response
type hall effect sensors to accomplish this sensing.
The high cut-off frequency of the filter will result in more susceptible and sensitive nuisance overcurrent trips. As shown in the previous test data, the oscillation frequency can become near 100kHz with 1000 feet motor lead length cable.
Location of the current sensor also affects the short circuit protection sensitivity. The stray capacitance associated with the flywheel diode and IGBT, and the stray inductance associated with the wire or the printed circuit foil used inside the inverter easily form a high order resonance circuits which sometimes amplify the high frequency oscillated current waveforms on the phase output back to the DC bus current, when the DC bus current sensors are used. Since the DC bus current is, by nature, the switching frequency based waveforms as shown in Figure 13, it is more difficult to deal with the filtering.
The transient surge on the induction motor due to the PWM switching can cause motor failure. This failure is primarily due to the insufficient insulation between the phases of motor three phase windings.
Field installations have experienced such failures not only at the terminals of the motor but also in the slots of the motor stator. In one case, 12 motors with 8 poles ranging between 0.5HP to 1.5HP failed when supplied by the PWM drive system. These failures can mostly occur inside the stator slots where concentric windings are used without phase insulation. In order to resolve this problem, winding layouts were changed to 6 pole without different windings sharing the same slot and phase insulation such as DMD(Dacron-Mylar-Dacron)or NOMEX can be used to provide additional protection to the motor leads. With these proper measures the motors are proven to be capable of withstanding the overvoltages due to the PWM switching.
So far both motor and inverter issues have been addressed.
For the motor, the motor must have a phase insulation to protect the overvoltage surge as well as the high dv/dt. Motors specifically designed for "inverter duty" are commercially available and implore these materials to protect insulation systems. In general, motors should have proper phase insulation which sustains a minimum of two times the DC bus voltage.
For the inverter, there are several recommendations:
To protect the inverter without nuisance overcurrent trips caused by high frequency ringing a key design function is placement of the current sensors. The following is a summary of these trade offs.
DC bus current sensor vs. output current sensor
Current sensing of either the DC bus current or the output current can be used to protect the IGBTs from faults. Each type of sensing has its own advantages and disadvantages. The main disadvantage of sensing the current on the DC bus is that high frequency ringing is present from the tank circuit formed from the inductance on the DC bus wiring and the capacitance across the IGBTs. In larger drives this ringing is at a frequency low enough that it adversely effects the sensing time, from the filtering on the output of the current sensor that is required to suppress this ringing. The main disadvantage of sensing the output current, is that a shoot-through current (when a lower and upper IGBT are on at the same time) can not be sensed. The result of these two problems is that usually DC bus current sensing can be used in smaller drives, and output current sensing plus desaturation detection is used in larger drives.
Drive system installation also has an effect. The following precautions can help in reducing high frequency current oscillation.
During the entire design process of locating drives and motors, lead length must be considered. IGBT based PWM drives offer the engineer alternatives when selecting equipment with motor lead length the primary issue. This paper presents IGBT characteristics, modulation design alternatives, analysis based on transmission line theory, Inverter/Motor effects, and installation precaution possibilities when selecting the proper variable speed drive for long motor lead length applications. If the horsepower is greater than 10HP in the 460VAC inverters, there will be no significant effect on both inverter and motor due to long lead cable length. 460 VAC IGBT PWM inverters below 10HP will create harmful high frequency oscillation on the voltage and current waveforms.
Solutions to long lead length applications include:
 John D. Kraus, Electromagnetics, 4th edition, McGraw-Hill, Inc, 1992.
 BINDER Andreas,"Armature Insulation Stress of Low Voltage A.C. Motors due to Inverter Supply",ICEM '94 Conference Record,vol-2,p.431.
 D.Potoradi, C.Hofmeier, R.Nuscheler,"Transient Overvoltages Caused by Switching of High Voltage Synchronous Machines and their Distribution in Stator Windings", ICEM '94 Conference Record, vol-2, p.644.